Decorrelator for upmixing systems

ABSTRACT

An improved decorrelator is disclosed that processes an input audio signal in two separate paths. In one path, a banded phase-flip filter is applied to lower frequencies of the input audio signal. In a second path, a frequency-dependent delay is applied to higher frequencies of the input audio signal. Signals from the two paths are combined to obtain an output signal that is psychoacoustically decorrelated with the input audio signal. The decorrelated signal can be mixed with the input audio signal without generating audible artifacts.

TECHNICAL FIELD

The present invention relates to decorrelation techniques that may beused to improve the performance of so-called “upmixing” devices thatgenerate multiple audio signals from a set of fewer audio signals.

BACKGROUND ART

Techniques for generating multiple audio signals from a set of feweraudio signals have been developed for many years and are used in avariety of upmixing devices such as the Dolby Pro Logic II decoderdescribed in Gundry, “A New Active Matrix Decoder for Surround Sound,”19th AES Conference, May 2001. The perceived performance of the upmixingdevices can generally be improved by decorrelation because at least somedegree of decorrelation in the upmixed signals generally increases theperceived width of the aural image achieved by playback of the upmixedsignals. Decorrelation can be obtained in a variety of known waysincluding simple delays and more complicated all-pass lattice filters.

Many conventional upmixing devices use one or more matrix structures toderive a number M output audio signals from a number N input audiosignals, where N is less than M. Some devices use active or variablematrix structures that are adapted in response to control signalsderived from the input audio signals. When decorrelation is used, anactive matrix structure is sometimes divided into two stages. The firststage derives 2M intermediate signals from the N input audio signals andthe second stage derives the M output audio signals from the 2Mintermediate signals. A decorrelation technique is applied to half ofthe 2M intermediate signals. The second stage generates output audiosignals with varying degrees of correlation by mixing amounts ofnon-decorrelated and decorrelated signals that are adapted in responseto the control signals.

The choice of decorrelation technique can have a profound effect on theperformance of an upmixing device. The inventors have determined thatthe performance of an upmixing device can be improved significantly ifthe decorrelation technique can satisfy three requirementssimultaneously: provide a decorrelated signal that does not soundsignificantly different from the non-decorrelated signal, provide asufficient amount of decorrelation to ensure the decorrelated signalsounds discrete or distinct with respect to the non-decorrelated signal,and allow mixing of the decorrelated signal and the non-decorrelatedsignal without generating audible artifacts. An additional advantage ofsuch a technique is that the upmixed signals can be downmixed to a fewernumber of input audio signals without generating objectionableartifacts.

DISCLOSURE OF INVENTION

It is an object of the present invention to provide forpsychoacoustically decorrelated signals that do not sound distorted,have a sufficient amount of decorrelation to ensure thepsychoacoustically decorrelated signals sound discrete or distinct withrespect to the input audio signals, and allow mixing of thepsychoacoustically decorrelated signals and non-decorrelated signalswithout generating audible artifacts.

The present invention is directed toward achieving a type ofdecorrelation that is referred herein as psychoacoustical decorrelation,which is related to but differs from conventional numerical correlation.The numerical correlation of two signals can be calculated using avariety of known numerical algorithms. These algorithms yield a measureof numerical correlation called a correlation coefficient that variesbetween negative one and positive one. A correlation coefficient with amagnitude equal to or close to one indicates the two signals are closelyrelated. A correlation coefficient with a magnitude equal to or close tozero indicates the two signals are generally independent of each other.

Psychoacoustical correlation refers to correlation properties of audiosignals that exist across frequency subbands that have a so-calledcritical bandwidth. The frequency-resolving power of the human auditorysystem varies with frequency throughout the audio spectrum. The humanear can discern spectral components closer together in frequency atlower frequencies below about 500 Hz but not as close together as thefrequency progresses upward to the limits of audibility. The width ofthis frequency resolution is referred to as a critical bandwidth and, asjust explained, it varies with frequency.

Two signals are psychoacoustically decorrelated if the average numericalcorrelation coefficient across a critical bandwidth is equal to or closeto zero. The correlation coefficient need not be equal to or close tozero at all frequencies but, if it does have a magnitude that departssignificantly from zero at some frequencies, the numerical correlationmust vary in such a way that the average numerical correlationcoefficient in a critical bandwidth is equal to or close to zero.

The object stated above is achieved by the invention as set forth in theindependent claims. Advantageous implementations are set forth in thedependent claims.

Features of the present invention and its preferred implementations maybe better understood by referring to the following discussion and theaccompanying drawings. The contents of the following discussion and thedrawings are set forth as examples only and should not be understood torepresent limitations upon the scope of the present invention.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a schematic block diagram of an exemplary upmixing device.

FIG. 2 is a schematic block diagram of a decorrelator.

FIG. 3 is graphical illustration of the impulse response of an exemplaryHilbert transform.

FIG. 4 is a graphical illustration of the imaginary part of a complexfrequency response of an exemplary Hilbert transform.

FIG. 5 is a graphical illustration of the impulse response of anexemplary sparse Hilbert transform.

FIG. 6 is a graphical illustration of the imaginary part of a complexfrequency response of an exemplary sparse Hilbert transform.

FIG. 7 is a graphical illustration of a frequency-domain magnituderesponse of an exemplary truncated sparse Hilbert transform.

FIG. 8 is a graphical illustration of the imaginary part of a complexfrequency response of an exemplary phase-flipping filter.

FIG. 9 is a graphical illustration of the impulse response of anexemplary phase-flipping filter.

FIG. 10 is a schematic block diagram of a device that may be used toimplement various aspects of the present invention.

MODES FOR CARRYING OUT THE INVENTION A. Introduction

FIG. 1 is a schematic block diagram of one upmixing device 10 thatincorporates various aspects of the present invention. The device 10receives N input audio signals and upmixes them into M output audiosignals, where M>N. In the example shown in the figure, N=2 and M=5. Thestage-1 matrix 12 generates 2M intermediate signals in response to the Ninput audio signals. The decorrelator 20 processes one half of the 2Mintermediate signals to generate M decorrelated intermediate signals,and the stage-2 matrix generates M output audio signals in response tothe M decorrelated intermediate signals and the M non-decorrelatedintermediate signals. When the decorrelator 20 is implemented accordingto teachings of the present invention, it provides psychoacousticallydecorrelated signals that do not sound significantly different from thenon-decorrelated input signals, it provides a sufficient amount ofpsychoacoustical decorrelation to ensure the decorrelated signals sounddiscrete or distinct with respect to the non-decorrelated input signals,and it allows mixing of the decorrelated signals and thenon-decorrelated input signals without generating audible artifacts. Thecontroller 11 generates control signals in response to the N input audiosignals that are used to adapt the operation of the stage-1 matrix 12and the stage-2 matrix 14. Additional information about theimplementation and adaptation of these matrices may be obtained frominternational patent application no. PCT/US 2005/030453 entitled“Multichannel Decorrelation in Spatial Audio Coding” published 9 Mar.2006 as publication no. WO 2006/026452 A1, and J. Breebaart et al.,“MPEG Spatial Audio Coding/MPEG Surround Overview and Current Status,”AES 119th Convention, New York, October 2005.

FIG. 2 is a schematic block diagram of one implementation of a portionof the decorrelator 20 that processes one of the intermediate signals.An input intermediate signal is passed along two differentsignal-processing paths. The lower-frequency path includes a phase-flipfilter 21 and a low pass filter 22. The higher-frequency path includes afrequency-dependent delay 23, a high pass filter 24 and a delaycomponent 25. The outputs of the delay 25 and the low pass filter 22 arecombined in the summing node 26. The output of the summing node 26 is adecorrelated intermediate signal that is psychoacoustically decorrelatedwith respect to the input intermediate signal.

The cut off frequencies of the low pass filter 22 and the high passfilter 24 should be chosen so that there is no gap between the passbandsof the two filters and so that the spectral energy of their combinedoutputs in the region near the crossover frequency where the passbandsoverlap is substantially equal to the spectral energy of the inputintermediate signal in this region. The amount of delay imposed by thedelay 25 should be set so that the propagation delay of thehigher-frequency and lower-frequency signal processing paths areapproximately equal at the crossover frequency.

The decorrelator 20 may be implemented in different ways. Even theexemplary implementation shown in the figure may be modified. Forexample, either one or both of the low pass filter 22 and the high passfilter 24 may precede the phase-flip filter 21 and thefrequency-dependent delay 23, respectively. The delay 25 may beimplemented by one or more delay components placed in the signalprocessing paths as desired.

The illustrated implementations of the decorrelator 20 electricallycombines the signals from the two signal-processing paths; however,these signals may be combined in other ways. In one alternativeimplementation, the two signals are combined acoustically. This may bedone by omitting the summing node 26 from the device 20 and processingthe signals from the higher-frequency and lower-frequency signalprocessing paths separately in the stage-2 matrix 24. The stage-2 matrix24 can generate a lower-frequency band signal and higher-frequency bandsignal for each of its M output audio signals to drive differentacoustic transducers, which allows these signals to be combinedacoustically.

B. Lower-Frequency Processing Path 1. Banded Phase-Flip Filter

An ideal implementation of the phase-flip filter 21 has a magnituderesponse of unity and a phase response that alternates or flips betweenpositive ninety degrees and negative ninety degrees at the edges of twoor more frequency bands within the passband of the filter. This bandedphase flip filter 21 may be viewed as an extension of the Hilberttransform. The impulse response of the Hilbert transform is shown in thefollowing equation and illustrated in FIG. 3:

$\begin{matrix}{{H(k)} = \left\{ \begin{matrix}{{2/k}\;\pi} & \left\{ {{odd}\mspace{14mu} k} \right\} \\0 & \left\{ {{even}\mspace{14mu} k} \right\}\end{matrix} \right.} & (1)\end{matrix}$Because the impulse response of the Hilbert transform is anodd-symmetric response, the frequency response of the transform is acomplex function of frequency that is purely imaginary. This frequencyresponse, expressed as a function of normalized frequency f/Fs, where Fsis the sample frequency, is illustrated in FIG. 4. When a Hilberttransform is applied to a signal, it imparts a negative ninety degreephase shift to positive frequencies and a positive ninety degree phaseshift to negative frequencies. Although the phase-flip filter 21 couldbe implemented by the Hilbert transform, this implementation would notbe satisfactory because its decorrelated output signal does not sounddiscrete or distinct with respect to the audio signal that is input tothe transform.

This deficiency may be overcome by implementing the phase-flip filter 12with a sparse Hilbert transform that has the impulse response shown inthe following equation:

$\begin{matrix}{{H_{s}(k)} = \left\{ \begin{matrix}{{2/k^{\prime}}\;\pi} & \left\{ {{{odd}\mspace{14mu} k^{\prime}} = {k/S}} \right\} \\0 & \left\{ {otherwise} \right\}\end{matrix} \right.} & (2)\end{matrix}$The impulse response of the sparse Hilbert transform, with S=6, isillustrated in FIG. 5. This impulse response also is an odd-symmetricresponse; therefore, the frequency response of this sparse transform isa complex function that is purely imaginary. The frequency response isillustrated in FIG. 6. The phase response flips between positive andnegative ninety degrees several times. The interval between adjacentflips is equal to Fs/2S.

When implemented by a sparse Hilbert transform, the phase-flip filter 21provides a decorrelated signal that generally does not sound distorted,has a sufficient amount of decorrelation to ensure it sounds discrete ordistinct with respect to the input signal, and can be mixed with theinput signal without generating audible artifacts. For practicalimplementations, however, the impulse response of the sparse Hilberttransform must be truncated. The length of the truncated response can beselected to optimize decorrelator performance by balancing a tradeoffbetween transient performance and smoothness of the frequency response.

On one hand, the impulse response should be short enough to provide goodtransient performance. If the impulse response is too long, transientswill be audibly smeared in the decorrelated output signal.

On the other hand, the impulse response should be long enough to providea reasonably smooth magnitude for its frequency response. FIG. 7illustrates the frequency-domain magnitude response of a sparse Hilberttransform with S=6 and a truncated impulse response with six non-zerocoefficients. The magnitude response contains notches at thosefrequencies where the phase flips occur. The width of these notches isinversely related to the length of the impulse response of the sparseHilbert transform. The notches become narrower as the impulse responseis lengthened. If the notches are too wide, the phase-flip filter 21will generate annoying artifacts in its decorrelated output signal.

The number of phase flips is controlled by the value of the S parameter.This parameter should be chosen to balance a tradeoff between the degreeof decorrelation and the impulse response length. A longer impulseresponse is required as the S parameter value increases. If the Sparameter value is too small, the filter provides insufficientdecorrelation. If the S parameter is too large, the filter will smeartransient sounds over an interval of time sufficiently long to createobjectionable artifacts in the decorrelated signal as discussed above.

The ability to balance these characteristics can be improved byimplementing the phase-flip filter 21 to have a non-uniform spacing infrequency between adjacent phase flips, with a narrower spacing at lowerfrequencies and a wider spacing at higher frequencies. Thisimplementation can provide on one hand narrower notches in thefrequency-domain magnitude response and more time smearing at lowerfrequencies, and can provide on the other hand wider notches in thefrequency-domain magnitude response and less time smearing at higherfrequencies. This implementation is preferred because it has been foundthat the effects of time smearing is less noticeable at low frequenciesand more noticeable at high frequencies, and the effects ofwidely-spaced notches are more noticeable at low frequencies but lessnoticeable at high frequencies.

In a preferred implementation of the phase-flip filter 21, the spacingbetween adjacent phase flips is a logarithmic function of frequency. Oneexample is illustrated in FIG. 8. The corresponding impulse response isillustrated in FIG. 9. This filter can be implemented as a finiteimpulse response (FIR) filter with an impulse response obtained by: (1)generating a function such as that shown in FIG. 8 with smoothinterpolations for the transitions between the function values ofpositive one and negative one; (2) creating a complex-valued frequencyresponse having a real part equal to zero and an imaginary part equal tothe function generated in the first step; and (3) applying an inverseFourier transform to the complex-valued frequency response to generatethe impulse response. Preferably, the filter is implemented by fastconvolution.

A notch exists in the frequency response for each transition in thephase response. The preferred implementation has a frequency responsewith notches having widths that are the greater of approximately 20 Hzor one-tenth an octave.

The phase-flip response may be illustrated by a complex-valued phasorthat is aligned with the imaginary axis and flips between oneorientation along the positive imaginary axis and a second orientationalong the negative imaginary axis. The phasor passes through zero whenit flips between orientations, which indicates the filter gain is zeroat these instants. This accounts for the notches in the frequencyresponse.

An alternative implementation can use a different phasor trajectory thatfollows the unit circle. This describes the frequency response of anall-pass filter. This filter can be implemented as an FIR filter with animpulse response obtained by: (1) generating a function such as thatshown in FIG. 8 with smooth interpolations for the transitions betweenthe function values of positive one and negative one; (2) creating acomplex-valued frequency response with a magnitude equal to one and aphase response in degrees equal to the function generated in the firststep multiplied by ninety so that the phase makes transitions betweenpositive ninety and negative ninety degrees; and (3) applying an inverseFourier transform to the complex-valued frequency response to generatethe impulse response. Preferably, the filter is implemented by fastconvolution.

The important characteristic of this as well as any other implementationof the phase-flip filter 21 is that the resulting filter has a bimodaldistribution in frequency of its phase response with peaks substantiallyequal to positive and negative ninety degrees. A peak is said to besubstantially equal to some nominal angle if it is within ten degrees.The frequency interval of the transitions between these two valuesshould be relatively small, and the frequency interval between adjacenttransitions should be small compared to the passband of the filter.

This FIR filter and the Hilbert transform filters discussed above arenot causal. In a practical implementation, the non-causal property isachieved with the use of a delay. This delay should be accounted for inthe higher-frequency path to keep the signals in these two paths alignedin time so that they can be combined properly by the summing node 26.The non-causal delay should also be accounted for in signal paths thatdo not pass through the decorrelator 20.

2. Low Pass Filter

The phase-flip filter 21 provides good decorrelation performance ofaudio signals up to approximately 2.5 kHz. Another mechanism that isdiscussed below is used for higher frequencies. A frequency limit can beimposed on the phase-flip filter 21 in a variety of ways including theuse of a low pass filter applied to its output, a low pass filterapplied to its input, or a modified design that incorporates the desiredlow-pass characteristic in the phase-flip filter itself. Conventionallinear filter design techniques may be used to obtain the modifieddesign.

C. Higher-Frequency Processing Path 1. Frequency-Dependent Delay

A process that delays an input signal and combines the delayed signalwith the non-delayed input signal operates like a comb-filter thatgenerates an output signal with notches in its spectrum. These notchesproduce annoying distortions in the combined output signal. Thefrequency dependent delay 23 avoids this problem by imposing a delaythat decreases with increasing frequency. The frequency-dependent delayproduces a non-uniform spacing between adjacent notches in the spectrumof the combined output signal, which can reduce the audibility ofartifacts produced by these notches for higher frequencies.

The frequency dependent delay 23 may be implemented by a filter that hasan impulse response equal to a finite length sinusoidal sequence h[n]whose instantaneous frequency decreases monotonically from π to zeroover the duration of the sequence. This sequence may be expressed as:h[n]=G√{square root over (|ω′(n)|)} cos(φ(n)),for 0≦n<L  (3)where

ω(n)=the instantaneous frequency;

ω′(n)=the first derivative of the instantaneous frequency;

G=normalization factor;

φ(n)=∫₀ ^(n)ω(t) dt=instantaneous phase; and

L=length of the delay filter.

The normalization factor G is set to a value such that:

$\begin{matrix}{{\sum\limits_{n = 0}^{L - 1}{h^{2}\lbrack n\rbrack}} = 1} & (4)\end{matrix}$

A filter with this impulse response can sometimes generate “chirping”artifacts when it is applied to audio signals with transients. Thiseffect can be reduced by adding a noise-like term to the instantaneousphase term as shown in the following equation:h[n]=G√{square root over (|ω′(n)|)} cos(φ(n)+N(n)),for 0≦n<L  (5)If the noise-like term is a white Gaussian noise sequence with avariance that is a small fraction of π, the artifacts that are generatedby filtering transients will sound more like noise rather than chirpsand the desired relationship between delay and frequency is stillachieved.

2. High Pass Filter

The frequency dependent delay 23 provides good decorrelation performanceof audio signals for frequencies above approximately 2.5 kHz. Afrequency limit can be imposed on the frequency dependent delay 23 in avariety of ways including the use of a high pass filter applied to itsoutput, a high pass filter applied to its input, or a modified designthat incorporates the desired high-pass characteristic in the frequencydependent delay filter itself. Conventional linear filter designtechniques may be used to obtain the modified design.

3. Delay

It is anticipated that in some implementations the group delay of thephase-flip filter 21 will exceed the minimum delay of the frequencydelay 23 at the highest frequency of interest. The delay 25 is providedin the higher-frequency path to account for the excess delay so that thesignals in the two paths can be combined to provide a decorrelatedsignal across the frequency band of interest. This delay can be insertedanywhere in the higher-frequency path. Alternatively, the frequencydependent delay 23 can be designed to provide the appropriate amount ofdelay.

D. Implementation

Devices that perform the processes for the processing paths may bedesigned in a variety of ways including discrete components for eachprocess, an FIR filter for each of the processing paths, and a singlecomposite FIR filter. The impulse response for this composite filter maybe obtained by implementing each processing path as a separatetime-domain to frequency-domain transform, combining thefrequency-domain responses of the two transforms, and obtaining theimpulse response of the composite filter by applying a frequency-domainto time-domain transform to the combined frequency-domain responses.

These devices may be implemented in a variety of ways including softwarefor execution by a computer or some other device that includes morespecialized components such as digital signal processor (DSP) circuitrycoupled to components similar to those found in a general-purposecomputer. FIG. 10 is a schematic block diagram of a device 70 that maybe used to implement aspects of the present invention. The DSP 72provides computing resources. Random access memory (RAM) 73 is used bythe DSP 72 for processing. ROM 74 represents some form of persistentstorage such as read only memory (ROM) for storing programs needed tooperate the device 70 and possibly for carrying out various aspects ofthe present invention. Input/output (I/O control 75 represents interfacecircuitry to receive and transmit signals by way of the communicationchannels 76, 77. In the embodiment shown, all major system componentsconnect to the bus 71, which may represent more than one physical orlogical bus; however, a bus architecture is not required to implementthe present invention.

In embodiments implemented by a general purpose computer system,additional components may be included for interfacing to devices such asa keyboard or mouse and a display, and for controlling a storage device78 having a storage medium such as magnetic tape or disk, or an opticalmedium. The storage medium may be used to record programs ofinstructions for operating systems, utilities and applications, and mayinclude programs that implement various aspects of the presentinvention.

These devices may also be implemented by discrete logic components,integrated circuits, one or more ASICs and/or program-controlledprocessors. The manner in which these devices are implemented is notimportant to the present invention.

Software implementations of the present invention may be conveyed by avariety of machine readable media such as baseband or modulatedcommunication paths throughout the spectrum including from supersonic toultraviolet frequencies, or storage media that convey information usingessentially any recording technology including magnetic tape, cards ordisk, optical cards or disc, and detectable markings on media includingpaper.

The invention claimed is:
 1. A method for decorrelating an input audiosignal that comprises: filtering the input audio signal according to afirst impulse response in a first frequency subband to generate a firstsubband signal that represents the input audio signal in the firstfrequency subband with a frequency-dependent change in phase having abimodal distribution in frequency with peaks substantially equal topositive and negative ninety-degrees, and according to a second impulseresponse in a second frequency subband to generate a second subbandsignal that represents the input audio signal in the second frequencysubband with a frequency-dependent delay, wherein: the second impulseresponse is not equal to the first impulse response, the secondfrequency subband includes frequencies that are higher than frequenciesincluded in the first frequency subband, the first frequency subbandincludes frequencies that are lower than frequencies included in thesecond frequency subband; the first impulse response represents a bandedphase-flip filter in cascade with a low-pass filter; and the secondimpulse response represents a frequency-dependent delay in cascade witha high-pass filter; and generating an output signal that represents acombination of the first subband signal and the second subband signal,and has a measure of mathematical correlation with the input audiosignal that varies over frequency and has averages across perceptualsubbands that are closer to zero than averages across narrowerbandwidths.
 2. The method of claim 1, wherein the high-pass filter andthe low-pass filter each have a cutoff frequency within the range from 1kHz to 5 kHz.
 3. The method of claim 1, wherein the second impulseresponse comprises a finite-length sinusoidal sequence.
 4. The method ofclaim 1, wherein the frequency-dependent change in phase has transitionsbetween positive and negative changes in phase at a plurality offrequencies within the second frequency subband.
 5. The method of claim4, wherein the transitions are separated by frequency intervals having awidth that is substantially equal to 150 Hz or 0.415 octave, whicheveris greater.
 6. An apparatus for decorrelating an input audio signal thatcomprises: means for filtering the input audio signal according to afirst impulse response in a first frequency subband to generate a firstsubband signal that represents the input audio signal in the firstfrequency subband with a frequency-dependent change in phase having abimodal distribution in frequency with peaks substantially equal topositive and negative ninety-degrees, and according to a second impulseresponse in a second frequency subband to generate a second subbandsignal that represents the input audio signal in the second frequencysubband with a frequency-dependent delay, wherein: the second impulseresponse is not equal to the first impulse response, the secondfrequency subband includes frequencies that are higher than frequenciesincluded in the first frequency subband, and the first frequency subbandincludes frequencies that are lower than frequencies included in thesecond frequency subband; the first impulse response represents a bandedphase-flip filter in cascade with a low-pass filter; and the secondimpulse response represents a frequency-dependent delay in cascade witha high-pass filter; means for generating an output signal thatrepresents a combination of the first subband signal and the secondsubband signal, and has a measure of mathematical correlation with theinput audio signal that varies over frequency and has averages acrossperceptual subbands that are closer to zero than averages acrossnarrower bandwidths.
 7. The apparatus of claim 6, wherein the high-passfilter and the low-pass filter each have a cutoff frequency within therange from 1 kHz to 5 kHz.
 8. The apparatus of claim 6, wherein thesecond impulse response comprises a finite-length sinusoidal sequence.9. The apparatus of claim 6, wherein the frequency-dependent change inphase has transitions between positive and negative changes in phase ata plurality of frequencies within the second frequency subband.
 10. Theapparatus of claim 9, wherein the transitions are separated by frequencyintervals having a width that is substantially equal to 150 Hz or 0.415octave, whichever is greater.
 11. A non-transitory medium recording aprogram of instructions that is executable by a device to perform amethod for decorrelating an input audio signal, wherein the methodcomprises: filtering the input audio signal according to a first impulseresponse in a first frequency subband to generate a first subband signalthat represents the input audio signal in the first frequency subbandwith a frequency-dependent change in phase having a bimodal distributionin frequency with peaks substantially equal to positive and negativeninety-degrees, and according to a second impulse response in a secondfrequency subband to generate a second subband signal that representsthe input audio signal in the second frequency subband with afrequency-dependent delay, wherein: the second impulse response is notequal to the first impulse response, the second frequency subbandincludes frequencies that are higher than frequencies included in thefirst frequency subband, and the first frequency subband includesfrequencies that are lower than frequencies included in the secondfrequency subband; the first impulse response represents a bandedphase-flip filter in cascade with a low-pass filter; and the secondimpulse response represents a frequency-dependent delay in cascade witha high-pass filter; generating an output signal that represents acombination of the first subband signal and the second subband signal,and has a measure of mathematical correlation with the input audiosignal that varies over frequency and has averages across perceptualsubbands that are closer to zero than averages across narrowerbandwidths.
 12. The non-transitory medium of claim 11, wherein thehigh-pass filter and the low-pass filter each have a cutoff frequencywithin the range from 1 kHz to 5 kHz.
 13. The non-transitory medium ofclaim 11, wherein the second impulse response comprises a finite-lengthsinusoidal sequence.
 14. The non-transitory medium of claim 11, whereinthe frequency-dependent change in phase has transitions between positiveand negative changes in phase at a plurality of frequencies within thesecond frequency subband.
 15. The non-transitory medium of claim 14,wherein the transitions are separated by frequency intervals having awidth that is substantially equal to 150 Hz or 0.415 octave, whicheveris greater.